Some Basics on Battery Ratings and Their Validation

A key aspect of optimizing battery run-time on battery powered mobile devices is measuring and analyzing their current drain to gain greater insight on how the device is making use of its battery power and then how to make better use of it. I went into a bit of detail on this in a previous posting, “Using Current Drain Measurements to Optimize Battery Run-time of Mobile Devices”.

A second aspect of optimizing battery run-time is making certain you are making optimum use of the battery powering the device. This starts with understanding and validating the battery’s stated capacity and energy ratings. Simply assuming the battery meets or exceeds its stated ratings without validating them is bound to leave you coming up shorter than expected on run-time.  It is critical that you validate them per the manufacturer’s recommended conditions. This serves as a starting point of finding out what you can ultimately expect from the battery you intend to use in your device. More than likely constraints imposed by the nature of your device and its operating conditions and requirements will further reduce the amount of capacity you can expect from the battery in actual use.

A battery’s capacity rating is the total amount of charge the battery can deliver. It is product of the current it can deliver over time, stated as ampere-hours (Ah) or miiliampere-hours (mAh). Alternately the charge rating is also stated as coulombs (C), where:

·         1 coulomb (C)= 1 ampere-sec

·         1 ampere-hour (Ah)= 3,600 coulombs

A battery’s energy rating is the total amount of energy the battery can deliver. It is the product of the power it can deliver over time, stated as watt-hours (Wh) or milliwatt-hours (mWh). It is also the product of the battery’s capacity (Ah) and voltage (V). Alternately the energy rating is also stated as joules (J) where:

·         1 joule (J)= 1 watt-second

·         1 watt-hour (Wh) = 3,600 joules

One more fundamental parameter relating to a battery’s capacity and energy ratings is the C rate or charge (or discharge) rate. This is the ratio of the level of current being furnished (or drawn from, when discharging) the battery, to the battery’s capacity, where:

·         C rate (C) = current (A) / (capacity (Ah)

·         C rate (C) = 1 / charge or discharge time

It is interesting to note while “C” is used to designate units of C rate, the units are actually 1/h or h^-1. The type of battery and its design has a large impact on the battery’s C rate. Batteries for power tools have a high C rate capability of 10C or greater, for example, as they need to deliver high levels of power over short periods of time. More often however is that many batteries used in portable wireless mobile devices need to run for considerably longer and they utilize batteries having relatively low C rates. A battery’s capacity is validated with a C rate considerably lower than it is capable of as when the C rate is increased the capacity drops due to losses within the battery itself.

Validating a battery’s capacity and energy ratings requires logging the battery’s voltage and current over an extended period of time, most often with a regulated constant current load. An example of this for a lithium ion cell is shown in Figure 1 below. Capacity was found to be 12% lower than its rating.

Figure 1: Measuring a battery’s capacity and energy

Additional details on this can be found in a technical overview I wrote, titled “Simply Validating a Battery’s Capacity and Energy Ratings”. As always, proper safety precautions always be observed when working with batteries and cells. Validating the battery’s stated capacity and energy ratings is the first step. As the battery is impacted by the device it is powering, it must then be validated under its end-use conditions as well. Stay tuned!

Battery drain analysis of handheld HP 973A multimeter

I have owned a Hewlett-Packard 973A multimeter for longer than I can remember. What has always amazed me about this meter is that I have never had to change the batteries in it! It runs off of 2 AA batteries (in series, of course), and earlier this week, I had to open it up to change a blown fuse for the mA/uA current measurement input (that’s what I get for lending the meter to someone).

While I had it open, I took a look at the AA batteries and was surprised to see a date code of 04-99. That means these batteries have been powering this multimeter for at least 13 years! I admit that I don’t use the meter very frequently, but I am still impressed with how long these batteries lasted. The series combination measured about 2.6 V – still plenty of charge left to power the multimeter (2 new batteries in series measure about 3.2 V).

Since we make power supplies that can perform battery drain analysis, I decided to take a quick look at the current drawn by the multimeter from these batteries. I used an Agilent N6705B mainframe with an N6781A Source/Measure Unit (SMU) installed. This SMU has many features that make it easy to analyze current drain. For example, I set the SMU for Current Measure mode which means it acts like a zero-burden ammeter (an ammeter with no voltage drop across the inputs). I found that the multimeter (set to measure DC V) draws about 3.5 mA from the 2.6 V series combination of AA batteries. I used both the Meter View feature of the SMU and the Data Logger to verify the current. The Data Logger shows the dynamic current being drawn from the batteries and I measured the average current between the markers.

Typical AA batteries are rated for about 2500 mA-hours, so with a 3.5 mA load, they will last more than 700 hours. It is no wonder that the batteries lasted a long time; I use this meter only a few hours per month, so assuming 3 hours per month, the batteries would last about 20 years!

While I had the back cover off, I removed the batteries and powered the multimeter directly from the N6781A SMU. I could then slowly lower the voltage and find when the low battery indicator came on. This happened at about 2.3 V. Continuing to lower the voltage, the LCD display continued to work down to almost 1.0 V. I also noticed that the current drawn by the multimeter increased as the voltage decreased – the multimeter was drawing a nearly constant amount of power from the source – roughly 9 to 10 mW.

I figured while I had the multimeter open, I might as well install new batteries. I doubt I will write another post the next time these batteries need to be replaced in 20 to 30 years, but keep checking here.… you never know!

*OPC and You

Today I have a guest-blogger, one of my Agilent colleagues (and friends), Matt Carolan. Matt has experience with programming our power products, so I asked him if he had anything he wanted to share with our audience. He has many things to talk about and will most likely contribute to future posts, but decided to start with the *OPC command, which is the “Operation Complete” command. Here is Matt’s post:

Hi, my name is Matt and I am an Application Support Engineer at Agilent Technologies. I have had 12 years of experience programming our power products and I wanted to write about a small but powerful command, *OPC.

*OPC is a standard IEEE-488.2 command that allows you to synchronize your power supply with your program. *OPC lets you know when all pending operations are complete. A pending operation is something such as the voltage being set or the output turning on. I worked as a test engineer for a few years and we always used the *OPC command in our calibration routines. We would send a calibration command (such as CALibration:LEVel:P1) followed by a *OPC? query. This allowed us to ensure that the calibration command had finished executing and that the power supply should be outputting the correct level before we took any measurements with our test system.

    

There are two ways to use *OPC. There is a standard *OPC command and the *OPC? query. The *OPC command will set bit 0 of the Standard Event Status register when all pending operations are complete. You can then use a *ESR? Command to poll the Standard Event Status Register. When this returns a 1, all pending operations are complete. When you use this command, it only works for any pending commands that were sent BEFORE you sent the *OPC command. It will not work for commands sent after the *OPC command. You can send another *OPC command to start the cycle again.

The other way is to use it as a query. When you send a *OPC? query, it will put a 1 in the output buffer when all pending operations are complete. The main drawback of this method is that if there is an operation that takes a long time to complete, your *OPC? query will timeout. You would need to have a long timeout set in your IO library to avoid this. You cannot send any commands after the query without getting an error so this will hold up your program until all pending operations are complete.

If you have any questions please comment here or on the Agilent forum at: https://www.agilent.com/find/forums

Battery-killing cell phone apps?

Two days ago, I came across an article entitled “Do Android Security Apps Kill Your Batteries?” The article talks about mobile device users avoiding security apps because they think the apps run down their batteries too quickly. A member of the Anti-Malware Testing Standards Organization (AMTSO) is using Agilent’s N6705B DC Power Analyzer to evaluate just how much the security apps affect battery run time. While the results are not yet complete, the researchers are planning to measure power usage with no security app running, with the app running in the background, and with the app actively working. Their full report is due out by the end of July. Here is a link to the article, written by Neil Rubenking in his SecurityWatch blog for PC Magazine Digital Edition:

https://securitywatch.pcmag.com/mobile-security/298170-do-android-security-apps-kill-your-batteries

I was pleased to see the N6705B DC Power Analyzer used in this way – this product has power modules and software that are specifically designed to do exactly this type of evaluation!

If you have to evaluate a mobile device’s battery run time for any reason, here is a link to “10 Tips to Optimize a Mobile Device’s Battery Life” written by our own Ed Brorein (contributor extraordinaire to this blog):

https://cp.literature.agilent.com/litweb/pdf/5991-0160EN.pdf

And here is a link to Ed’s post from a few months ago on “Using Current Drain Measurements to Optimize Battery Run-time of Mobile Devices”:

https://powersupplyblog.tm.agilent.com/2012/03/using-current-drain-measurements-to.html

When the researchers complete and publish their evaluation on how security apps affect your cell phone battery run time, we’ll be sure to follow-up with another post! In the mean time, protect your phone in whatever way you like, and keep charging ahead by charging your batteries!

What Is Old is New Again: Soft-Switching and Synchronous Rectification in Vintage Automobile Radios

I have to admit I am a bit of a vintage electronics technologist.  One of many pass times includes bringing vintage vacuum tube automobile radios back to life. In working with modern DC sources I’ve seen innovations come about in the past decade for efficient power conversion, including soft switching and synchronous rectification. A funny thing however, for those who have been around long enough, or into vintage technologies like me, is that these issues and somewhat comparable solutions existed up to 70 years ago for automobile radios and other related electronic equipment. What is old is new again!

As we know, vacuum tubes (or valves to many) were to electronics back then as what semiconductors are to electronics today. The problem for portable and mobile equipment was that the vacuum tubes needed typically 100 or more volts DC to operate. They did have high voltage batteries for portable equipment but for automobiles the radio really needed to run off the 6 or 12 volts DC available from the electrical system. The solution: A DC/DC boost converter!

Up until the mid 1950’s most all automobile radios used vacuum tubes biased with high voltage generated from a rather primitive but clever DC/DC boost converter design. The inherent technological challenge was semiconductors did not yet exist to chop up the low-voltage, high-current DC to convert it to high-voltage, low-current DC. Of course if the semiconductors did exist this would all be a moot point! Making use of what was available the DC/DC boost converters employed what were called vibrators, which are a form of a continuously buzzing relay, to chop up the low-voltage DC for conversion. Maybe some of you are familiar with the soft humming sound heard when an original vintage automobile radio is turned on, prior to the vacuum tubes finally warming up and the audio taking over? That humming is the vibrator, the “heart” of the DC/DC boost converter in the radio.

Figure 1 below is an example circuit of vibrator-based DC/DC boost converter in a vintage automobile radio. This is just one of quite variety of different implementations created back then. Two pairs of contacts in the vibrator act in a push-pull fashion to convert the low-voltage DC into a low-voltage AC square wave. This in turn is converted to a high-voltage square wave by the transformer. Because the vibrator is an electro-mechanical device, it is limited in how fast it can switch. Switching frequencies are typically about 100 to 120 Hz. The transformers used are naturally the steel-laminated affairs similar in nature to the transformers used to convert household line voltage in home appliances. Very possibly some radio manufacturers used off- the-shelf appliance transformers in reverse to step up the voltage!  Often a small rectifier vacuum tube, such as a 6X4 (relatively modern, by vacuum tube standards) would be used to convert the high voltage AC to high voltage DC, but in this particular example I am showing here another two pairs of contacts on the secondary side switch simultaneously with the first pairs of contacts to rectify the high voltage AC. Highly efficient synchronous rectification, up to 70 years ago!

Figure 1: Representative DC/DC boost converter for a vintage automobile radio

Establishing Measurement Integration Time for Leakage Currents

The proliferation of mobile wireless devices drives a corresponding demand for components going into these devices. A key attribute of these components is the need to have low levels of leakage current during off and standby mode operation, to extend the battery run-time of the host device. I brought up the importance of making accurate leakage currents quickly in an earlier posting “Pay Attention to the Impact of the Bypass Capacitor on Leakage Current Value and Test Time”(click here to review). Another key aspect about making accurate leakage currents quickly is establishing the proper minimum required measurement integration time. I will go into factors that govern establishing this time here.

Assuming the leakage current being drawn by the DUT, as well as any bypass capacitors on the fixture, have fully stabilized, the key thing with selecting the correct measurement integration time is getting an acceptable level of measurement repeatability. Some experimentation is useful in determining the minimum required amount of time. The primary problem with leakage current measurement is one of AC noise sources present in the test set up. With DC leakage current being just a few micro amps or less these noises are significant. Higher level currents can be usually measured much more quickly as the AC noises are relatively negligible in comparison. There are a variety of potential noise sources, including radiated and conducted from external sources, including the AC line, and internal noise sources, such as the AC ripple voltage from the DC source’s output. This is illustrated in Figure 1 below. Noise currents directly add to the DC leakage current while noise voltages become corresponding noise currents related by the DUT and test fixture load impedance.

Figure 1: Some noise sources affecting DUT current measurement time

Using a longer measurement time integrates out the peak-to-peak random deviations in the DC leakage current to provide a consistently more repeatable DC measurement result, but at the expense of increasing overall device test time. Measurement repeatability should be based on a statistical confidence level, which I will do into more detail further on. Using a measurement integration time of exactly one power line cycle (1 PLC) of 20 milliseconds (for 50 Hz) or 16.7 milliseconds (for 60 Hz) cancels out AC line frequency noises. Many times a default time of 100 milliseconds is used as it is an integer multiple of both 20 and 16.7 milliseconds. This is fine if overall DUT test time is relatively long but generally not acceptable when total test time is just a couple of seconds, as is the case with most components. As a minimum, setting the measurement integration time to 1 PLC is usually the prudent thing to do when short overall DUT test time is paramount.

Reducing leakage current test time below 1 PLC means reducing any AC line frequency noises to a sufficiently low level such that they are relatively negligible compared to higher frequency noises, like possibly the DC source’s wideband output ripple noise voltage and current. Proper grounding, shielding, and cancellation techniques can greatly reduce noise pickup. Paying attention to the choice and size of bypass capacitors used on the test fixture is also important. A larger-than-necessary bypass capacitor can increase measured noise current when the measuring is taking place before the capacitor, which is many times the case. Establishing the requirement minimum integration time is done by setting a setting an acceptable statistical confidence level and then running a trial with a large number of measurements plotted in a histogram to assure that they fall within this confidence level for a given measurement integration time. If they did not then the measurement integration time would need to be increased. As an example I ran a series of trials to determine what the acceptable minimum required integration time was for achieving 10% repeatability with 95% confidence for a 2 micro amp leakage current. AC line noises were relatively negligible. As shown in Figure 2, when a large series of measurements were taken and plotted in a histogram, 95% of the values fell within +/- 9.5% of the mean for a measurement integration time of 1.06 milliseconds.

Figure 2: 2 Leakage current measurement repeatability histogram example

Leakage current measurements by nature take longer to measure due to their extremely low levels. Careful attention to minimizing noise and establishing the minimum required measurement integration time contributes toward improving the test throughput of components that take just seconds to test.

Can a standard DC power supply be used as a current source?

The quick answer to this question is, yes, most standard DC power supplies can be used as current sources. However, this question deserves more attention, so what follows is the longer answer.

Most DC power supplies can operate in constant voltage (CV) or constant current (CC) mode. CV mode means the power supply is regulating the output voltage and the output current is determined by the load connected across the output terminals. CC mode means the power supply is regulating the output current and the output voltage is determined by the load connected across the output terminals. When operating in CC mode, the power supply is acting like a current source. So any power supply that can operate in CC mode can be used as a current source (click here for more info about CV/CC operation).

Is a standard power supply a good current source?
An ideal current source would have infinite output impedance (an ideal voltage source would have zero output impedance). No power supply has infinite output impedance (or zero output impedance) regardless of the mode in which it is operating. In fact, most power supply designs are optimized for CV mode since most power supply applications require a constant voltage. The optimization includes putting an output capacitor across the output terminals of the power supply to help lower output voltage noise and also to lower the output impedance with frequency. So the effectiveness of a standard power supply as a current source will depend on your needs with frequency.

At DC, a power supply in CC mode does make a good current source. Typical CC load regulation specifications support this notion (click here for more info about load regulation). For example, an Agilent N6752A power supply (maximum ratings of 50 V, 10 A, 100 W) has a CC load regulation specification of 2 mA. This means that the output current will change by less than 2 mA for any load voltage change. So when operating in CC mode, a 50 V output load change will produce a current change of less than 2 mA. If we take the delta V over worst case delta I, we have 50 V / 2 mA = 25 kΩ. This means that the DC output impedance will always be 25 kΩ or more for this power supply. In fact, the current will likely change much less than 2 mA with a 50 V load change making the DC output impedance in CC mode much greater than 25 kΩ.

Of course, a power supply’s effectiveness as a current source should be judged by the output impedance beyond the DC impedance. See the figure below for a graph of the N6752A CC output impedance with frequency:

If the graph continued in the low frequency direction, the output impedance would continue to rise as a “good” current source should. At higher frequencies, the CC loop gain inside the product begins to fall. As the loop gain moves through unity and beyond, the output capacitor in the supply dominates the behavior of the output impedance, so at high frequencies, the output impedance is lower. So how good the power supply is as a current source depends on your needs with frequency. The higher the output impedance, the better the current source. The output impedance also correlates to the CC transient response (and to a much lesser extent, the output programming response time).

The bottom line here is that in most applications, a standard DC power supply can be used in CC mode as a current source.

Experiences with Power Supply Common Mode Noise Current Measurements

I wrote an earlier posting “DC Power Supply Common Mode Noise Current Considerations” (click here to review) as common mode noise current can be an issue in electronic test applications we face. This is not so much of an issue with all-linear based power supply designs as it can be for ones incorporating switching based topologies. High performance DC power supplies designed for test applications should have relatively low common mode current by design. I thought this would be a good opportunity to get some more first-hand experience validating common mode noise current. The exercise proved to be a bit of an eye-opener. I tried different approaches and, no surprise; I got back seemingly conflicting results. Murphy was busy working overtime here!

I settled on a high performance, switching-based DC source on having a low common mode noise characteristic of 10 mA p-p and 1 mA RMS over a 20 Hz to 20 MHz measurement bandwidth. To properly make this measurement the general consensus here is a wide band current probe and oscilloscope is the preferred solution for peak to peak noise, and a wide band current probe and wide band RMS voltmeter is the preferred solution for RMS noise. As the wide band RMS voltmeters are pretty scarce here I relied on the oscilloscope for both values for the time being. The advantages of current probes for this testing are they provide isolation and have very low insertion impedance.

I located group’s trusty active current probe and oscilloscope. The low signal level I intended to measure dictated using the most sensitive range providing 10 mA/div (with oscilloscope set to 10 mV/div).
One area of difficulty to anticipate with modern digital oscilloscopes is there are a lot of acquisition settings to contend with, all having a major impact on the actual reading. After sorting all of these out I finally got a base line reading with my DC source turned off, shown in Figure 1.

Figure 1: Common mode noise current base-line reading

My base-line reading presented a bit of a problem. With 1 mV corresponding to 1 mA my 2.5 mA p-p / 0.782 mA RMS base-line values were a bit high in comparison to my expected target values. It would be nicer for this noise floor to be at about 10X smaller so that I don’t have to really factor it out. Resorting to the old trick of looping the wire through the current probe 5 times gave me a 5X larger signal without changing the base-line noise floor. The oscilloscope was now displaying 2 mA /div, with 1 mV corresponding to 0.2 mA. In other words my base-line is now 0.5 mA p-p / 0.156 mA RMS. The penalty for doing this is of course more insertion impedance. Now I was all set to measure the actual common mode noise current. Figure 2 shows the common mode noise current measurement with the DC source on.

Figure 2: Common mode noise current measurement

Things to pay attention to include checking the current on both + and – leads individually to earth ground and load the output with an isolated load (i.e. a power resistor). Full load most often brings on worst case values. Based on the 0.2 conversion ratio I’m now seeing 8 mA p-p and 1.12 mA RMS, including the baseline noise. I am reasonably in the range of the expected values and having a credible measurement!

I decided to compare this approach to making a 50 ohm terminated direct connection. This set up is depicted in Figure 3 below.

Figure 3: 50 ohm terminated directly connected common mode noise current measurement

I knew insertion impedance was considerably more with this approach so I tried both 10 ohm and 100 ohm shunt values to see what kind of readings I would end up with. Table 1 summarizes the results for the directly connected measurement approach.

Table 1: 50 ohm terminated directly connected common mode noise current results

Clearly the common mode noise current results were nowhere near what I obtained with using a current probe, being much lower, and also highly dependent on the shunt resistor value. Why is that? Looking more closely at the results, the voltage values are relatively constant for both shunt resistor cases. Beyond a certain level of increasing shunt resistance the common mode noise behaves more as a voltage than a current. For this particular DC source the common mode voltage level is extremely low, just a few millivolts.

Not entirely content with the results I was getting I located a different high performance DC source that also incorporated switching topology. No actual specifications or supplemental characteristics had been given for it. When tested it exhibited considerably higher common mode noise than the first DC source. The results are shown in table 2 below.

Table 2: 50 ohm terminated directly connected common mode noise current results, 2nd DC source

With both voltage and current results changing for these two test conditions the common mode noise is exhibiting somewhere between being a noise current versus being a noise voltage. I had hoped to see what the results would be using the current probe but it seemed to have walked away when I needed it!

In Summary:
Making good common mode current noise measurements requires paying a lot of attention to the choice of equipment, equipment settings, test set up, and DUT operating conditions. I still have bit more to investigate but at least I have a much better understanding as to what matters. Maybe in a future posting I can provide what could be deemed as the “golden set up”! To get results that correlate reasonably with any stated values will likely require a set up that exhibits minimal insertion impedance across the entire frequency spectrum. Making directly coupled measurements without the use of a current probe will prove challenging except maybe for DC sources having rather high levels of common mode noise currents

The underlying concern here of course is what is what will be the impact to the DUT due to any common mode noise current from the test system’s DC source. Generally that is any common mode noise current ends up becoming differential mode noise voltage on the DUT’s power input due to impedance imbalances. But one thing I found from my testing is that the common mode noise is not purely a current with relatively unlimited compliance voltage but somewhere between being a noise voltage and noise current, depending on loading conditions. For the first DC source, with what appears to be only a few millivolts behind the current it is unlikely that it would create any issues for even the most sensitive DUTs. For the second DC source however, having 100’s of millivolts behind its current, could potentially lead to unwanted differential voltage noise on the DUT. Further investigation is in order!

Pay Attention to the Impact of the Bypass Capacitor on Leakage Current Value and Test Time

It is no secret there is big demand for all kinds of wireless battery powered devices and, likewise, the components that go into these devices. These devices and their components need to be very efficient in order to get the most operating and standby time out of the limited amount of power they have available from the battery. Off-mode and leakage currents of these devices and components need to be kept to a minimum as an important part of maximizing battery run and standby time. Levels are typically in the range of tens of microamps for devices and just a microamp or less for a component. Off-mode and leakage currents are routinely tested in production to assure they meet specified requirements. The markets for wireless battery powered devices and their components are intensely competitive. Test times need to be kept to a minimum, especially for the components. It turns out the choice of the input power bypass capacitor being used, either within the DUT on the DUT’s test fixture, can have a large impact on the leakage current value and especially the test time for making an accurate leakage current measurement.

Good things come in small packages?
A lot has been done to provide greater capacitance in smaller packages for ceramic and electrolytic capacitors, for use in bypass applications. It is worth noting that electrolytic and ceramic capacitors exhibit appreciable dielectric absorption, or DA. This is a non-linear behavior causing the capacitor to have a large time-dependent charge or discharge factor, when a voltage or short is applied. It is usually modeled as a number of different value series R-C pairs connected in parallel with the main capacitor. This causes the capacitor to take considerable time to reach its final steady state near-zero current when a voltage is applied or changed. When trying to test the true leakage current on a DUT it may be necessary to wait until the current on any bypass capacitors has reached steady state before a current measurement is taken. Depending on the test time and capacitor being used this could result in an unacceptably long wait time.

So how do they compare?
In Figure 1 I captured the time-dependent current response waveform for a 5.1 megohm resistor, a 5.1 megohm resistor in parallel with 100 microfarad electrolytic capacitor, and finally a 5.1 megohm resistor in parallel with 100 microfarad film capacitor, when a 5 volt step stimulus was applied.

Figure 1: Current response of different R-C loads to 5 volt step

The 5.1 megohm resistor (i.e. “no capacitor”) serves as a base line to compare the affect the two different bypass capacitors have on leakage current measurement. The film capacitor has relatively ideal electrical characteristics in comparison to an equivalent electrolytic or ceramic capacitor. It settles down to near steady state conditions within 0.5 to 1 second. At 3 to 3.5 seconds out (marker placement in Figure 1) the film capacitor is contributing a fairly negligible 42 nanoamps of additional leakage. In comparison the electrolytic capacitor current is still four times as great as the resistor current and nowhere near being settled out. If you ever wondered why audio equipment producers insist on high performance film capacitors in critical applications, DA is one of those reasons!

So how long did it take for the electrolytic capacitor to reach steady state? I set up a longer term capture in Figure 2 for the electrolytic capacitor. After about a whopping 40 seconds later it seemed to be fully settled out, but still contributing a substantial 893 nanoamps of additional steady state leakage current.

Figure 2: 100 microfarad electrolytic capacitor settling time

Where do I go from here?
So what should one do when needing to test leakage current? When testing a wireless device be aware of what kind and value of bypass capacitor has been incorporated into it. Most likely it is a ceramic capacitor nowadays. Film capacitors are too large and cost prohibitive here. Find out how long it takes to settle to its steady state value. Also, off-state current measurements are generally left until the end of the testing to not waste time waiting for the capacitor to reach steady state. If testing a component, if a bypass capacitor is being used on the test fixture, consider using a film capacitor. With test times of just seconds and microamp level leakage currents the wrong bypass capacitor can be a huge problem!

What’s the difference between a fixed output power supply and a programmable output power supply?

I am very fortunate to work with a lot of very smart, talented, and knowledgeable engineers with vast technical backgrounds. I also work with some very smart, talented, and knowledgeable non-technical individuals, some of whom are involved in our sales process. Last month, during a sales training session, one of these individuals identified a competitor’s power supply product that looked very similar to one of our Agilent power supply products: a mainframe with plug-in modules. Upon further investigation, it turned out that the competitor’s product really consisted of modules that were virtually fixed output power supplies while our Agilent product provides programmable output power supplies. So, in fact, these two products do not compete against each other despite the initial appearance. This experience inspired me to post about the differences between a fixed output DC power supply and a programmable output DC power supply.

Fixed output power supplies
A fixed output power supply has, well, a fixed output voltage. This means that when the power supply is plugged in and the output is on, the output voltage is a single voltage that is not expected to change – it is fixed at that voltage. These power supplies are typically used to provide simple bias for a circuit. Some are embedded on a printed circuit board or mounted inside a larger chassis with other circuits, and others may be rack mounted. Fixed output power supplies come in many forms as shown below. Some have a single output voltage while others provide multiple output voltages. One example of a fixed output power supply with multiple outputs is a PC supply (upper left in the figure) – it typically has the following DC output voltages: +3.3 V, +5 V, and +/- 12 V. These voltages provide power to the chips on the PC’s motherboard, including the microprocessor, and to the peripherals installed in the PC, such as the disk drive.

Fixed output power supplies normally have a fixed current limit setting. They typically regulate their output voltages to an accuracy of a few percent (for example, 5%). Many have output noise specifications of 50 to 150 mV peak-to-peak and typically have no measurement capability (such as output voltage or output current measurement).

Programmable output power supplies
A programmable output power supply’s output voltage can be set (programmed) by the user. This means that you can set the voltage to any value between zero and the maximum rated voltage (plus and/or minus) of the supply and change it whenever necessary. The set values are normally controlled either from the front panel of the supply with knobs or buttons, or through the built-in interface connected to a computer. Commands are sent from the computer to the supply to change its output voltage. These power supplies are typically used in test and measurement applications. They might be found on a design engineer’s bench or mounted in a rack of automated test equipment. They come in many forms as shown below. Some have a single output voltage while others provide multiple output voltages. The ability to change the output voltage is required in a circuit test environment. For example, to test a PC’s disk drive, you will need +5 V and + 12 V to power the drive. When installed in a PC, the disk drive will get power from a fixed output power supply in the PC. But when testing the disk drive outside of the PC, you should use a programmable power supply. Since the output voltage of a fixed output supply has an accuracy of a few percent, the voltage could be higher or lower than the nominal. For example, if the +5 V fixed supply has an accuracy of 5%, it could be any value from +4.75 V to +5.25 V. When installed in the PC, the disk drive has to work over this entire range of possible voltages applied to it. So to test it outside of the PC, a programmable power supply should be used and set to various voltages in this range to ensure the drive will always work.


Programmable output power supplies normally have a programmable current limit setting to help protect the device under test from exposure to excessive current. They typically regulate their output voltages to an accuracy of a few tenths of a percent or even better (for example, 0.06%). They have output noise specifications of 1 to 50 mV peak-to-peak and typically have built-in output voltage and output current measurement capability.

Summary
So the main differences between fixed output power supplies and programmable output power supplies are the ability to change the output voltage and the specifications. You can change the output voltage of a programmable supply while that of a fixed supply cannot be changed. Programmable supplies have much more accurate output voltages and much lower noise. They also can typically measure their own output voltage and current while a fixed output supply cannot. Of course, the extra capabilities of the programmable supplies add to their price, but you get what you pay for!

Protect your DUT with power supply features including a watchdog timer

The two biggest threats of damage to your device under test (DUT) from a power supply perspective are excessive voltage and excessive current. There are various protection features built into quality power supplies that will protect your DUT from exposure to these destructive forces. There are also some other not-so-common features that can prove to be invaluable in certain applications.

Soft limits
The first line of defense against too much voltage or current can be using soft limits (when available). These are maximum values for voltage and current you can set that later prevent someone from setting output voltage or current values that exceed your soft limit settings. If someone attempts to set a higher value (either from the front panel or over the programming interface), the power supply will ignore the request and generate an error. While this feature is useful to prevent accidentally setting voltages or currents that are too high, it cannot protect the DUT if the voltage or current actually exceeds a value due to another reason. Over-voltage protection and over-current protection must be used for these cases.

Over-voltage protection
Over-voltage protection (OVP) is a feature that uses an OVP setting (separate from the output voltage setting). If the actual output voltage reaches or exceeds the OVP setting, the power supply shuts down its output, protecting the DUT from excessive voltage. The figure below shows a power supply output voltage heading toward 20 V with an OVP setting of 15 V. The output shuts down when the voltage reaches 15 V.

Some power supplies have an SCR (silicon-controlled rectifier) across their output that gets turned on when the OVP trips essentially shorting the output as quickly as possible. Again, the idea here is to protect the DUT from excessive voltage by limiting the voltage magnitude and exposure time as much as possible. The SCR circuit is sometimes called a “crowbar” circuit since it acts like taking a large piece of metal, such as a crowbar, and placing it across the power supply output terminals.

Over-current protection
Over-current protection (OCP) is a feature that uses the constant current (CC) setting. If the actual output current reaches or exceeds the constant current setting causing the power supply to go into CC mode, the power supply shuts down its output, protecting the DUT from excessive current. The figure below shows a power supply output current heading toward 3 A with a CC setting of 1 A and OCP turned on. The power supply takes just a few hundred microseconds to register the over-current condition and then shut down the output. The CC and OCP circuits are not perfect, so you can see the current exceed the CC setting of 1 A, but it does so for only a brief time.

The OCP feature can be turned on or off and works in conjunction with the CC setting. The CC setting prevents the output current from exceeding the setting, but it does not shut down the output if the CC value is reached. If OCP is turned off and CC occurs, the power supply will continue producing current at the CC value basically forever. This could damage some DUTs as the undesired current flows continuously through the DUT. If OCP is turned on and CC occurs, the power supply will shut down its output, eliminating the current flowing to the DUT.

Note that there are times when briefly entering CC mode is expected and an OCP shutdown would be a problem. For example, if the load on the power supply has a large input capacitor, and the output voltage is set to go from zero to the programmed value, the cap will draw a large inrush current that could temporarily cause the power supply to go into CC mode while charging the cap. This short time in CC mode may be expected and considered acceptable, so there is another feature associated with the OCP setting that is a delay time. Upon a programmed voltage change (such as from zero to the programmed value as mentioned above), the OCP circuit will temporarily ignore the CC status just for the delay time, therefore avoiding nuisance OCP tripping.

Remote inhibit
Remote inhibit (or remote shutdown) is a feature that allows an external signal, such as a switch opening or closing, to shutdown the output of the power supply. This can be used for protection in a variety of ways. For example, you might wire this input to an emergency shutdown switch in your test system that an operator would use if a dangerous condition was observed such as smoke coming from your DUT. Or, the remote inhibit could be used to protect the test system operator by being connected to a micro switch on a safety cover for the DUT. If dangerous voltages are present on the DUT when operating, the micro switch could disable DUT power when the cover is open.

Watchdog timer
The watchdog timer is a unique feature on some Agilent power supplies, such as the N6700 series. This feature looks for any interface bus activity (LAN, GPIB, or USB) and if no bus activity is detected by the power supply for a time that you set, the power supply output shuts down. This feature was inspired by one of our customers testing new chip designs. The engineer was running long-term reliability testing including heating and cooling of the chips. These tests would run for weeks or even months. A computer program was used to control the N6700 power supplies that were responsible for heating and cooling the chips. If the program hung up, it was possible to burn up the chips. So the engineer expressed an interest in having the power supply shut down its own outputs if no commands were received by the power supply for a length of time indicating that the program has stopped working properly. The watchdog timer allows you to set delay times from 1 to 3600 seconds.

Other protection features that protect the power supply itself
There are some protection features that indirectly protect your DUT by protecting the power supply itself, such as over-temperature (OT) protection. If the power supply detects an internal temperature that exceeds a predetermined limit, it will shut down its output. The temperature may rise due to an unusually high ambient temperature, or perhaps due to a blocked or incapacitated cooling fan. Shutting down the output in response to high temperature will prevent other power supply components from failing that could lead to a more catastrophic condition.

One other way in which a power supply protects itself is with an internal reverse protection diode across its output terminals. As part of the internal design, there is often a polarized electrolytic capacitor across the output terminals of a power supply. If a reverse voltage from an external power source was applied across the output terminals, the cap (or other internal circuitry) could easily be damaged. The design includes a diode across the output terminals with its cathode connected to the positive terminal and its anode connected to the negative terminal. The diode will conduct if a reverse voltage from an external source is applied across the output terminals, thereby preventing the reverse voltage from rising above a diode drop and damaging other internal components.

What Is Going On When My Power Supply Displays “UNR”?

Most everyone is familiar with the very traditional Constant Voltage (CV) and Constant Current (CC) operating modes incorporated in most any lab bench or system power supply. All but the most very basic power supplies provide display indicators or annunciators to indicate whether it is in CV or CC mode. However, moderately more sophisticated power supplies provide additional indicators or annunciators to provide increased insight and more information about their operating status. One annunciator you may encounter is seeing “UNR” flash on, either momentarily or continuously. It’s fairly obvious that this means that the power supply is unregulated; it is failing to maintain a Constant Voltage or Constant Current. But what is really going on when the power supply displays UNR and what things might cause this?
To gain better insight about CV, CC and UNR operating modes it is helpful to visualize what is going on with an IV graph of the power supply output in combination with the load line of the external device being powered. I wrote a two part post about voltage and current levels and limits which you may find useful to review. If you like you can access it from these links levels and limits part 1 and levels and limits part 2. This posting builds nicely on these earlier postings. A conventional single quadrant power supply IV graph with resistive load line is depicted in Figure 1. As the load resistance varies from infinity to zero the power supply’s output goes through the full range of CV mode through CC mode operation. With a passive load like a resistor you are unlikely to encounter UNREG mode, unless perhaps something goes wrong in the power supply itself.
Figure 1: Single quadrant power supply IV characteristic with a resistive load

However, with active load devices you have a pretty high chance of encountering UNR mode operation, depending where the actual voltage and current values end up at in comparison to the power supply’s voltage and current settings. One common application where UNR can be easily encountered is charging a battery (our external active load device) with a power supply. Two different scenarios are depicted in Figure 2. For scenario 1, when the battery voltage is less than the power supply’s output, the point where the power supply’s IV characteristic curve and the battery’s load line (a CV characteristic) intersect, the power supply is in CC mode, happily supplying a regulated charge current into the battery. However, for scenario 2 the battery’s voltage is greater than the power supply’s CV setting (for example, you have your automobile battery charger set to 6 volts when you connect it to a 12 volt battery). Providing the power supply is not able to sink current the battery forces the power supply’s output voltage up along the graph’s voltage axis to the battery’s voltage level. Operating along this whole range of voltage greater than the power supply’s output voltage setting puts the power supply into its UNR mode of operation.
Figure 2: Single quadrant power supply IV characteristic with a battery load

A danger here is more sophisticated power supplies usually incorporate Over Voltage Protection (OVP). One kind of OVP is a crowbar which is an SCR designed to short the output to quickly bring down the output voltage to protect the (possibly expensive) device being powered. When connected to a battery if an OVP crowbar is tripped, damage to the power supply or battery could occur due to batteries being able to deliver a fairly unlimited level of current. It is worth knowing what kind of OVP there is in a power supply before attempting to charge a battery with it. Better yet is to use a power supply or charger specifically designed to properly monitor and charge a given type of battery. The designers take these things into consideration so you don’t have to!
I have digressed here a little on yet another mode, OVP, but it’s all worth knowing when working with power supplies! Can you think of other scenarios that might drive a power supply into UNR? (Hint: How about the other end of the power supply IV characteristic, where it meets the horizontal current axis?)

If you need fast rise and fall times for your DUT power, use a power supply with a downprogrammer

If you have to provide DC power to a device under test (DUT) and you want the voltage fall time to be just as fast as the rise time, use a power supply with a downprogrammer. A downprogrammer is a circuit built into the output of a power supply that actively pulls the output voltage down when the power supply is moving from a higher setting to a lower setting. Power supplies are good at forcing their output voltage up since that is what their internal circuitry is designed to do. This design results in fast rise times. However, when the supply’s output is changed to move down in voltage, the power supply’s output capacitor (and any additional external DUT capacitance) will need to be discharged. Without a downprogrammer, if there is a light load or no load on the output of the power supply, there is nowhere for the current from the output cap to flow to discharge it. This scenario causes the voltage to take a long time to come down resulting in slow fall times. And this behavior leads to longer test times since you will have to wait for the output voltage to settle to the lower value before you can proceed with your test.

The figures below show an example of the output voltage rise and fall times of a power supply without a downprogrammer under light load conditions. You can see the short rise time (tens of milliseconds) and longer fall time (several seconds).



One of my colleagues, Bob Zollo, wrote an article on this topic that appeared in Electronic Design on February 7, 2012. Here is a link to the article:

https://electronicdesign.com/article/test-and-measurement/If-Your-Power-Supply-Needs-Fast-Rise-And-Fall-Times-Try-A-Down-Programmer-64725

A power supply without an active downprogrammer can have fall times that are tens to hundreds of times longer than a power supply with a downprogrammer. If your test requires you to have fast fall times for your DUT power, or your test requires you to frequently change the voltage on your DUT (both up and down) and throughput is an issue for you, make sure the power supply you choose has a downprogrammer – you won’t have to wait as long for the voltage to move from a higher value to a lower value.

Using Current Drain Measurements to Optimize Battery Run-time of Mobile Devices

One power-related application area I do a great deal of work on is current drain measurements and analysis for optimizing the battery run-time of mobile devices. In the past the most of the focus has been primarily mobile phones. Currently 3G, 4G and many other wireless technologies like ZigBee continue to make major inroads, spurring a plethora of new smart phones, wireless appliances, and all kinds of ubiquitous wireless sensors and devices. Regardless of whether the device is overly power-hungry due to running data-intensive applications or power-constrained due to its ubiquitous nature, there is a need to optimize its thirst for power in order to get the most run-time from its battery. The right kind of measurements and analysis on the device’s current drain can yield a lot of insight on the device’s operation and efficiency of its activities that are useful for the designer in optimizing its battery run-time. I recently completed an article that appeared in Test & Measurement World, on-line back in November and then in print in their Dec 2011- Jan 2012 issue. Here is a link to the article:
https://www.tmworld.com/article/520045-Measurements_optimize_battery_run_time.php

A key factor in getting current drain measurements to yield the deeper insights that really help optimize battery run-time is the dynamic range of measurement, both in amplitude and in time, and then having the ability to analyze the details of these measurements. The need for a great dynamic range of measurement stems from the power-savings nature of today’s wireless battery powered devices. For power-savings it is much more efficient for the device to operate in short bursts of activities, getting as much done as possible in the shortest period of time, and then go into a low power idle or sleep state for an extended period of time between these bursts of activities. Of course the challenge for the designer to get his device to quickly wake up, stabilize, do its thing, and then just as quickly go back to sleep again is no small feat! As one example the current drain of a wireless temperature transmitter for its power-savings type of operation is shown in Figure 1.


Figure 1: Wireless temperature transmitter power-savings current drain

The resulting current drain is pulsed. The amplitude scale has been increased to 20 µA/div to show details of the signal’s base. This particular device’s current drain has the following characteristics:
• Period of ~4 seconds
• Duty cycle of 0.17%
• Currents of 21.8 mA peak and 53.7 µA average for a crest factor of ~400
• Sleep current of 7 µA
This extremely wide dynamic range of amplitude is challenging to measure as it spans about 3 ½ decades. Both DC offset error and noise floors of the measurement equipment must be extremely low as to not limit needed accuracy and obscure details.

Likewise being able to examine details of the current drain during the bursts of activities provides insights about the duration and current drain level of specific operations within the burst. From this you can make determinations about efficiencies of the operations and if there is opportunity to further optimize them. As an example, in standby operation a mobile phone receives in short bursts about every 0.25 to 1 seconds to check for incoming pages and drops back into a sleep state in between the receive (RX) bursts. An expanded view of one of the RX current drain bursts is shown in figure 2.


Figure 2: GPRS mobile phone RX burst details

There are a number of activities taking place during the RX burst. Having sufficient measurement bandwidth and sampling time resolution down to 10’s of µsec provides the deeper insight needed for optimizing these activities. The basic time period for the mobile phone standby operation is on the order of a second but it is usually important to look at the current drain signal over an extended period of time due to variance of activities that can occur during each of the RX bursts. Having either a very deep memory, or even better, high speed data logging, provides the dynamic range in time to get 10’s of µsec of resolution over an extended period of time, so that you can determine overall average current drain while also being able to “count the coulombs” it takes for individual, minute operations, and optimize their efficiencies.

Anticipate seeing more here in future posts about mobile wireless battery-powered devices, as it relates to the “DC” end of the spectrum. In the meantime, while you are using your smart phone or tablet and battery life isn’t quite meeting your expectation (or maybe it is!), you should also marvel at how capable and compact your device is and how far it has come along in contrast to what was the state-of-the-art 5 and 10 years ago!

On DC Source Voltage and Current Levels and (Compliance) Limits Part 2: When levels and limits are not the same

In part 1 my colleague made a good argument for current and voltage level and limit settings actually being one and the same thing and it was really just a case of semantics whether your power supply was operating in constant voltage or in constant current mode. I disagreed and I was not ready to admit defeat on this yet. Now is my chance to explain why I believe they’re not one and the same thing.

I have been doing quite a bit of work with source measure units (SMUs) that support multi quadrant output operation. They in fact feature (constant) voltage sourcing and current sourcing modes of operation. This tailors the operation of the SMU for operating as a voltage source with a set current compliance range or conversely as a current source with a set voltage compliance range. Right at the start one difference is the set up conditions. The output voltage or current level is set to zero while the corresponding current or voltage limit is set to some value, often maximum, so that the DC source accordingly starts out in either constant voltage or constant current for normal operating conditions.

Some products feature a programmable or fixed power limits. In one product I know of, the programmable power limit acts accordingly to override and cut back the either the voltage limit when set for current sourcing, or the current limit when set for voltage sourcing. It does not do this in true real-time however. It cuts back the limit based on the level setting, as a convenient means as to help prevent the user from accidently over-powering the DUT. Alternately many auto-ranging output DC power sources exist that provide an extended range of output and voltage for a given output power capacity. They incorporate a fixed power limit to protect the power supply itself from being inadvertently overloaded, as shown in Figure 1. Usually the idea is for the user to stay below the limit, not operate in power limit. The point here on these examples is that the power parameter is an example of being a limit but not really a level.

Figure 1: Auto-ranging DC power supply power limit

More to the point is some SMUs may incorporate two limits to provide a bounded compliance range with specified positive and negative limits. Not all DUTs are passive, non-reactive devices. As one illustrative example a DUT may be the output of 2-quadrant DC voltage source which you want to force up or down, within limits, or a battery you want to charge and discharge at a fixed rate, with your test system DC source. This set up is illustrated in Figure 2.

Figure 2: Test system DC source driving the output of a DUT source

Figure 3 shows the constant voltage or voltage priority output characteristic for one particular SMU having two programmable current limits. Clearly both limits cannot also be the current level setting as you can only have one level setting. For the case of the external voltage source load line #1 (not all load lines are resistances!), when SMU voltage is less than the DUT source voltage (VEXT1 load line), the current is –ILIM. Conversely when SMU voltage is greater than the DUT source voltage (VEXT2 load line), the current is then +ILIM. In the case of the battery as a DUT this can be used to charge and discharge the battery to specified voltage levels. This desired behavior is achieved using voltage priority operation. Current priority operation would yield very different results. Understanding the nuances of voltage priority, current priority, levels, and limits is useful for getting more utility from your DC sources for more unusual and challenging power test challenges.

Figure 3: Example of a current priority output characteristic driving a DUT voltage source

In closing I’ll concur with my colleague, in many test situations using most DC sources the voltage and current levels and limits may not have a meaningful difference. However, in many more complex cases, especially when dealing with active DUTs and using more capable DC sources and SMUs, there is a clear need for voltage and current level and limit controls that are clearly differentiated and not one and the same! What do you believe?

On DC Source Voltage and Current Levels and (Compliance) Limits Part 1: When levels and limits are one and the same

I was having a discussion with a colleague about constant current operation versus constant voltage operation and the distinction between level settings and limit settings the other day. “The level and limit settings are really the same thing!” he claimed. I disagreed. We each then made ensuing arguments in defense of our positions.

He based his argument on the case of a DC power supply that has both constant voltage and constant current operation. I’ll agree that is a reasonable starting point. As a side note there is a general consensus here that if it isn’t a true, well regulated constant voltage or constant current, whether settable or fixed, then it is simply a limit, not a level setting, end of story. He continued “if the load on the power supply is such that it is operating in constant voltage, then the voltage setting is the level setting and the current setting is the limit setting. If the load increases such that the power supply changes over from constant voltage operation into constant current operation then the voltage setting is becomes the limit setting and the current setting becomes the level setting!” (See figure 1.) He certainly has a good point! For your more basic DC power supply that only operates in quadrant 1 capable of sourcing power only, the current and voltage settings usually interchangeably serve as both the level and compliance limit setting, depending on whether the DC power supply is operating in constant voltage or constant current. The level and compliance limit regulating circuits are one and the same. Likewise with the programming, there are only commands to set the voltage and current levels. There are not separate commands for the limits. I might be starting to lose grounds on this discussion!
Figure 1: Unipolar single quadrant DC source operation

However, all is not lost yet. The DC power supply world is often more complicated than just this unipolar single quadrant operation just presented. Watch for my second part on when the levels and limits are not necessarily one and the same.